Sensing device and method for measuring emission time delay during irradiation of targeted samples

ABSTRACT

An apparatus for measuring emission time delay during irradiation of targeted samples by utilizing digital signal processing to determine the emission phase shift caused by the sample is disclosed. The apparatus includes a source of electromagnetic radiation adapted to irradiate a target sample. A mechanism generates first and second digital input signals of known frequencies with a known phase relationship, and a device then converts the first and second digital input signals to analog sinusoidal signals. An element is provided to direct the first input signal to the electromagnetic radiation source to modulate the source by the frequency thereof to irradiate the target sample and generate a target sample emission. A device detects the target sample emission and produces a corresponding first output signal having a phase shift relative to the phase of the first input signal, the phase shift being caused by the irradiation time delay in the sample. A member produces a known phase shift in the second input signal to create a second output signal. A mechanism is then provided for converting each of the first and second analog output signals to digital signals. A mixer receives the first and second digital output signals and compares the signal phase relationship therebetween to produce a signal indicative of the change in phase relationship between the first and second output signals caused by the target sample emission. Finally, a feedback arrangement alters the phase of the second input signal based on the mixer signal to ultimately place the first and second output signals in quadrature. Mechanisms for enhancing this phase comparison and adjustment technique are also disclosed.

CONTRACTUAL ORIGIN OF THE INVENTION

This invention was made with U.S. Government support under contractNAS9-97080 awarded by NASA and contract F33615-97-0729 awarded by theDepartment of the Air Force. The Government has certain rights in thisinvention.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to sensing instruments and methods formeasuring the concentration of an analyte in a medium and, moreparticularly, to a device and method for measuring such concentrationsby measuring the emission time delay during irradiation of a targetedsample surrounded by the analyte. Specifically, the present inventionrelates to a device and method for measuring exponential time constants,phase shifts, time delays and parameters derivable therefrom caused byirradiation of a targeted sample utilizing digital signal processing andespecially luminescence quenching systems, phase shifts throughnetworks, and time delays of photon migration through media.

2. Description of the Prior Art

Dynamic phase modulation, quenched luminescence sensors are well known.Instruments of this type have been, for example, developed or proposedfor use in hospitals to monitor the concentration of gases such asoxygen, ionized hydrogen and carbon dioxide within the blood ofpatients. The particular substance of interest, for example oxygen, isknown as the analyte.

As is known in the art, luminescence materials absorb energy and aredriven from their ground state energy level to an excited state energylevel in response to the application of energy from an electromagneticradiation source such as light. These materials are unstable in theirexcited states, and they luminesce or give off excess energy as theyreturn to their ground state. For example, the short wavelengthultraviolet light of black light stimulates dyes in a colored fabric toemit longer wavelengths, such as blue, green or red, and thus fluoresce.For the purposes of the present disclosure, the term "luminescence" asused herein is a general term which describes both fluorescence andphosphorescence, for all three terms are frequently used interchangeablyin the art. The distinction and overlap of the terms is obvious to oneskilled in the art.

In the presence of certain chemicals, many fluorescent materials aresaid to be quenched, i.e. the time constant of the fluorescence emissionis altered by the effects of the surrounding chemicals. The degree ofquenching of the fluorescence in turn can be related to theconcentration of the quencher, which for example may be a chemicaldissolved in water or mixed in air, such as oxygen in the blood ofpatients as explained above. There is a substantial amount of literaturethat describes fluorescent molecules that are selectively quenched byoxygen, carbon dioxide, glucose, pH, NH₃, metal ions, temperature andother environmentally and medically important analytes. These analytesare relevant to applications such as monitoring drinking water quality,industrial process control, monitoring of human respiratory function,human blood analysis for critical care patients, and the like.

One of the obstacles to the commercialization of fluorescence sensingdevices has been a lack of inexpensive yet accurate instrumentation forthe measurement of changes in the fluorescent time constant. Forexample, U.S. Pat. Nos. 4,845,368, 5,257,202, 5,495,850, 5,515,864 and5,626,134 all disclose devices for measuring analyte concentrationlevels based on fluorescence. However, these particular devices aregenerally expensive and complicated.

The fluorescence lifetime or time constant, τ, is the amount of time ittakes the fluorescence emission to decrease by a factor of 1/e or about63% after termination of irradiation as disclosed in U.S. Pat. No.4,716,3632 by Dukes et al, in column 1, lines 37-41. This is commonknowledge and is available in the literature reference, i.e. "Topics inFluorescence Vol. 2--Principles", ed. Joseph Lakowicz. If lightmodulated sinusoidally at a frequency, f, is thus applied to thefluorescence sensor, the output is a sinusoidal emission of identicalfrequency, but having a phase shift and reduced amplitude with respectto the excitation signal. The equation governing the relationshipbetween modulation frequency, f, phase shift, θ, and the fluorescenttime constant, τ, is as follows: ##EQU1##

Thus, if we know the excitation modulation frequency and can measure thephase shift of the emission signal relative to the excitation signal, wecan determine the fluorescence constant, τ, using the above Equation 1.In a fluorescence-based sensor, the fluorescence time constant ismeasured since this fluorescence time constant is altered by thepresence of certain chemical species. Consequently, the concentration ofchemical species can be determined by measuring the fluorescence timeconstant by measuring the phase shift associated therewith.

According to Equation 1, in order to measure the fluorescence timeconstant, one must know the excitation modulation frequency, f, and thephase shift of the light through the fluorescence system. With thesequantities, the fluorescence time constant can be calculated and thenrelated to analyte concentration. There are several different knowntechniques for determining the excitation frequency and phase shift of asystem with an unknown time constant. One manner of determining this isby exciting the sample with a fixed frequency signal and then measuringthe phase shift that results, that is the sample excitation modulationfrequency is maintained constant while the signal phase, which varieswith analyte concentration, is measured. U.S. Pat. Nos. 5,317,162,5,462,879, 5,485,530 and 5,504,337 all disclose such fixed frequency,variable phase techniques and devices. Of particular interest is anarticle by Venkatesh Vadde and Vivek Srinivas entitled, "A closed loopscheme for phase-sensitive fluorometry", the American Institute ofPhysics, Rev. Sci. Instrum., Vol. 66, No. 7, July 1995, p. 3750.

Another principal way of conducting the above measurements is byexciting the sample with a modulation frequency that maintains aconstant phase relationship between the excitation signal and theemission signal, that is the excitation frequency is varied in order tomaintain a particular phase relationship. Such devices and techniquesare known as phase-modulation, fluorescence-based sensing devices andare clearly illustrated in U.S. Pat. Nos. 4,840,485, 5,196,709, and5,212,386, and in an article by Brett A. Feddersen, et al. entitled,"Digital parallel acquisition in frequency domain fluorimetry", AmericanInstitute of Physics, Rev. Sci. lnstrum., Vol. 60, No. 9, September1989, p. 2929. Of particular interest is U.S. Pat. No. 4,716,3632 byDukes et al., which describes a feedback system that provides themodulation frequency required to give a constant phase shift of about45°. The resulting frequency is then used to determine the analyteconcentration which is a function of excited state lifetime.

U.S. Pat. No. 5,818,582 teaches the use of a DSP for fluorescencelifetime measurements, though not using quadrature signal comparison fordetermination of fluorescent sample phase shifts.

Despite the availability of the above-discussed techniques and sensingdevices, there is a continuing need for improved fluorescence-basedsensing instruments. In particular, there is a need for such deviceswhich are useful for a broad range of applications involving exponentialdecay and time delay measurements, which are made from inexpensivecomponents, and which present measurements in real time without the needfor off-line signal processing as is the case of the patents toFederson, Gratton and others. A major detriment to many of the devicespresently available is that they are very expensive to acquire andmaintain. Moreover, analog systems of the present art are subject todrift and therefore unnecessary errors. Such systems should be, to thecontrary, inexpensive, convenient to use and provide adequatesensitivity over an extended and continuous measurement range. Thesystem of the Dukes patent emphasizes optimal sensitivity over a widemeasurement range, but in so doing, requires very complex and expensivesystem components. To the contrary, optimal sensitivity can besacrificed for sub-optimal, adequate sensitivity in order to achieveinexpensive, less complicated measurement techniques. In addition, themeasurement approach of such devices should be susceptible to convenientand precise readout.

SUMMARY OF THE INVENTION

Accordingly, it is one object of the present invention to provide anapparatus and method for measuring emission time delay duringirradiation of targeted samples.

It is another object of the present invention to provide sensinginstruments which are applicable to a broad range time delay, phaseshift and exponential decay measurements involving luminescent materialsand various scattering media.

Yet another object of the present invention is to providefluorescence-based sensing instruments which are made from inexpensivecomponents.

Still another object of the present invention is to provide an apparatusand method for measuring emission time delay during irradiation oftargeted samples utilizing digital signal processing to determine theemission phase shift caused by the sample.

A further object of the present invention is to provide an apparatus andmethod for measuring luminescence-quenching systems, specifically oxygensensitive systems.

To achieve the foregoing and other objects and in accordance with thepurpose of the present invention, as embodied and broadly describedherein, an apparatus is disclosed for measuring emission time delayduring irradiation of targeted samples by utilizing digital signalprocessing to determine the emission phase shift caused by the sample.The apparatus includes a source of electromagnetic radiation adapted toirradiate a target sample. A mechanism generates first and seconddigital input signals of known frequencies with a known variable phaserelationship, and a device then converts the first and second digitalinput signals to analog sinusoidal signals. An element is provided todirect the first input signal to the electromagnetic radiation source tomodulate the source by the frequency thereof to irradiate the targetsample and generate a target sample emission. A device detects thetarget sample emission and produces a corresponding first output signalhaving a phase shift relative to the phase of the first input signal,the phase shift being caused by the emission time delay in the sample. Amember produces a known phase shift in the second input signal to createa second output signal. A mechanism is then provided for converting eachof the first and second analog output signals to digital signals. Amixer receives the first and second digital output signals and comparesthe signal phase relationship therebetween to produce a signalindicative of the change in phase relationship between the first andsecond output signals caused by the target sample emission. Finally, afeedback arrangement alters the phase of the second input signal basedon the mixer signal to ultimately place the first and second outputsignals in quadrature. Mechanisms for enhancing this phase comparisonand adjustment technique are also disclosed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings which are incorporated in and form a part ofthe specification illustrate preferred embodiments of the presentinvention and, together with a description, serve to explain theprinciples of the invention. In the drawings:

FIG. 1 is a schematic illustrating an embodiment of the presentinvention utilizing a direct phase adjustment, constant frequencytechnique with digital signal processing for measuring emission phaseshift to determine time delay through an irradiated sample.

FIG. 2 is a schematic illustrating another embodiment of the presentinvention utilizing variable-frequency and variable-phase techniques formeasuring emission phase shift to determine time delay through anirradiated sample.

FIG. 3 is a schematic illustrating yet another embodiment of the presentinvention similar to that of FIG. 2 but incorporating signal downconversion steps.

FIG. 4 is a schematic illustrating yet another embodiment of the presentinvention similar to that of FIG. 3 but incorporating dual quadraturesignal down conversion steps.

FIG. 5 is a schematic illustrating yet another embodiment of the presentinvention similar to that of FIG. 4 but eliminating the means for thedownconverting of high frequency signals to lower frequencies forquadrature phase detection.

FIG. 6 is a schematic illustrating yet another embodiment of the presentinvention using a single analog timing element and a DSP for real timedetermination of phase and lifetime.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring initially to FIG. 1, a closed-loop lifetime measurement device10 is illustrated which incorporates a Digital Signal Processor (DSP)12. Venkatesh and Srinivas, in the prior art references discussed above,disclose a closed-loop fluorescent-decay time measurement system thatallows a phase demodulator to operate in an optimal null condition. Thisis accomplished using two analog timing elements including acrystal-controlled frequency generator and a second analog timingelement to vary the phase of a measurement reference signal so thatsignals to the demodulator are always in quadrature. An important aspectof this disclosed technique is that the reference channel is phaseadvanced in order to maintain quadrature at the phase demodulator.

Within the DSP 12 of FIG. 1, all the components referred to below aredigital and defined by software loaded into the DSP chip. In the DSP 12,a constant frequency, dual-output, variable-phase waveform generator 14is provided and adapted to generate a first experimental digital signal16 and a second reference digital signal 26. The digital signals 16, 26are preferably identical in frequency. The experimental signal 16 isdirected through a digital-to-analog converter 20 where it is convertedinto an analog signal which drives an electromagnetic radiation-emittingdevice 22. The dual-output, variable-phase waveform generator 14 causesthe reference signal 26 to be phase advanced a known number of degrees(N°) relative to experimental signal 16. Reference signal 26 is thendirected through a second digital-to-analog converter 28.

In preferred form, the device 22 is a light emitting diode (LED). Thedevice 22 is activated by the analog form of the signal 16 and generatesa light emission 30. The emission 30 is directed to a target sample 32which includes therein material which will emit energy 34 as a result ofbeing impinged by the light 30. In one preferred form of the invention,the target sample 32 is a fluorescent material designed to generate afluorescence emission 34 upon contact with the LED emission 30. However,it should be understood that the target sample 32 may include anyappropriate emission-delay generating system as discussed above.

The emissions 34 are detected by a device 36, which in preferred form isa photodiode. The detection device 36 then generates an output signal 38which has a frequency identical to the experimental signal 16 but isphase retarded as a result of the time delay imposed by the targetsample 32. Thus, the output signal 38 is now phase-shifted an unknownamount relative to the experimental signal 16 and the reference signal26. In preferred form, the output signal 38 is directed through apre-amplifier 40. A pair of anti-aliasing filters 42, 44 are provided,and the output signal 38 passes through the filter 42 while thereference signal 26 passes through the filter 44, thereby becomingoutput reference signal 46. Thus, both the experimental signal 16 andreference signal 26 are effectively treated substantially identicallyoutside the DSP 12 except for the phase shift resulting from the targetsample 32. This is due to the fact that while the LED 22, the photodiode36 and the pre-amplifier 40 all add phase shifts to the experimentalsignal 16 that do not exist in the reference signal 26 chain, theseshifts, though significant, are calibrated out. In the preferredembodiment, the phase shift caused by LED 22, the photodiode 36 and thepre-amplifier 40 is negligible compared to the phase shift caused byanti-aliasing filters 42 and 44. The system 10 uses duplicateanti-aliasing filters 42, 44 to eliminate major phase imbalances thatwould otherwise exist in the two channels. Both signals 38 and 46 arepassed through respective analog-to-digital converters 48, 50 to createcounterpart digital output signals within the DSP 12.

The output reference signal 46 is then mixed with the outputexperimental signal 38 at the signal mixing device 52, which in thisparticular embodiment is preferably a phase demodulator. As waspreviously stated, the original reference signal 26 is phase-shifted bythe digital waveform generator 14 a specific number of degrees so thatthe experimental input signal 16 and the reference input signal 26 areout-of-phase by a known, predetermined amount. However, due to the phaseshift imposed by the target sample 32, the phase differences between theoutput signals 38 and 46 at the mixing member 52 are unknown. Thereforeand in preferred form, the phase demodulator 52 indicates when thereference signal 46 (A) relative to the experimental signal 38 (B) arein quadrature, or 90° apart. In other words, 90°-B=A if the signals arein quadrature. If the signals 38 and 46 are in fact in quadrature, thenno changes are made to the relative phase offset between input signals16 and 26. However, due to the phase shift caused by the target sample32, the signals 38 and 46 are not initially in quadrature at 52.

As a result, the phase demodulator 52 generates a signal 54 comprised ofboth AC and DC components, the DC component represents the phasedifference between the signals 38 and 46 relative to 90°. This signal 54is preferably passed through a low pass filter 56 to remove the ACcomponent creating a DC error signal 58. The sign and magnitude of theDC error signal 58 indicates the relative phase difference between inputsignals 38 and 46 and is preferably zero when the input signals 38 and46 are in quadrature. Based on the error signal 58, the digital waveformgenerator 14 continuously modifies the phase advance (N°) of thereference signal 26. In this manner, the device 10 continues to changethe relative phase of the signals 16 and 26 until the phases of theoutput experimental signal 38 and the output reference signal 46 are inquadrature at the phase demodulator 52, at which point error signal 58is substantially zero. At this stage, the phase shift through the sample32 is 90°-N°, with N° being the phase advance of the signal 26 relativeto signal 16. This phase shift quantity is then utilized with the knownfrequency of the experimental signal 16 and Equation 1 to calculate thelifetime of the target sample 32, which in turn will provide the desiredinformation about the analyte surrounding the target sample 32 asdiscussed above.

One of the important aspects of the embodiment illustrated in FIG. 1 isthat this embodiment utilizes parallel analog paths for both thereference and experimental signals in combination with digitalprocessing. These parallel paths are used for two principal reasons. Thefirst is that the digital-to-analog converters 20, 28 as well as theanalog-to-digital converters 48, 50 introduce time delays into thesignals passing through them. Any difference in time delay between thetwo paths will result in an undesired phase-offset between them.However, if both of the experimental and reference signals 16, 26 passthrough matched identical converters 20, 28 and 48, 50, thereconstruction and digitization will not result in a relative time delayof one signal with respect to the other.

The second reason is that the anti-aliasing filter 42 introduces asignificant amount of phase lag into the experimental analog signal 38,i.e. about 26 degrees at 20 kHz. By passing the reference analog signal26 through an identical anti-aliasing filter 44, the phase lag as aresult of the anti-aliasing filter is canceled. Additionally, any driftin phase cause by the anti-aliasing filter 42 in the experimental signalpath 38 will tend to be canceled by similar drift in the anti-aliasingfilter in the reference signal path 26. The symmetrical treatment of theexperimental 38 and reference 46 signals means that the phase differencebetween them is due only to the phase delay resulting from the targetsample 32 as well as the known phase advance created by the digitalphase shifter 24. While the LED driver circuit, the LED, the photodiodeand the preamplifier all contribute small phase shifts in theexperimental signal 16 that are not cancelled by similar components inthe reference signal 26 path, these phase shifts are cancelled in otherways as indicated above. In addition, since all operations within theDSP 12 are digital calculations, they are free from any drift ornon-linearity whatsoever.

In the above embodiment of FIG. 1, the DSP 12 implementation of thedevice 10 utilizes only one analog timing element for the generation ofthe reference and experimental signals 26, 16, for the basic phaseshifting of the reference signal 26, and for the phase demodulation ofsignals 46 and 38 at the demodulator 52. The time base for the DSPfrequency generation and the phase shifting is preferably derived from asingle external crystal oscillator. Moreover, the phase shifting of thereference signal 26 is preferably accomplished by the addition of two32-bit numbers which does not introduce a phase jitter as is true ofpure analog systems or of systems using more than one analog timingelement for signal generation, comparison and phase shifting. As aresult of the lack of phase jitter or instability (drift) between thetwo signals 16, 26, extremely small phase changes, e.g. 0.001 degrees,caused by the target sample 32 are detectable by the device 10.Moreover, the advantage of using identical anti-aliasing filters 42, 44is that any changes in the filter properties resulting from changingtemperatures are reflected in both the experimental and reference signalpaths and are therefore canceled. This substantially reduces electronicphase drift as compared to prior art devices.

As was discussed above, existing devices also utilize changes infrequency, rather than phase, to measure fluorescence emissions or thelike. In particular, U.S. Pat. No. 4,716,363 by Dukes et al. describes afluorescence lifetime measurement system that operates in this manner.In particular, the excitation frequency of Dukes is varied such that aconstant predetermined phase shift is obtained through the fluorescenceexperiment. The predetermined phase shift is selected to achieve optimalsensitivity to changes in lifetime. Since the frequency is inverselyproportional to the lifetime as illustrated above in Equation 1, thefrequency can be directly related to the quencher or analyteconcentration of the target sample 32 thereby circumventing the need tocalculate lifetime or phase.

While this particular system of Dukes operates fine in certaininstances, there are significant drawbacks. Without going into adetailed discussion of this reference, the Dukes' system operates suchthat the oscillator frequency is adjusted to maintain a fixed andoptimum phase-shift through the fluorescence experiment. However, insensing applications where the change in the lifetime of the fluorescentexperiment is large, the frequency must change over an equally largerange. Thus, if the lifetime changes by a factor of 100, then theoscillator must change frequency by a factor of 100. The use of smalleror larger phase-offsets will shift the maximum and minimum frequenciesup and down but will not compress the required range. There are manysituations where generating frequencies over such a wide range isimpractical because it is prohibitively complex or expensive. As aresult of this problem, the present invention provides the additionalembodiments of the invention as illustrated in FIGS. 2-4.

Referring now in particular to FIG. 2, it should be understood that likecomponents throughout all of the Figures and embodiments of the presentinvention will have like numerals and indicators. In this particularembodiment of FIG. 2, the device 10 includes a DSP 60. The DSP 60preferably includes a dual-output, variable-phase waveform generator 62.Within the waveform generator 62 is a frequency and phase calculator 64which determines the appropriate frequency and phase relationship of thetwo signals 16 and 26 output by generator 62. The frequency generator 62generates an initial experimental signal 16 with a known phase and aninitial reference signal 26 which is phase advanced relative to theexperimental signal 16 by the calculator 64.

As with the prior embodiment of FIG. 1, the signals 16, 26 pass throughtheir respective digital-to-analog converters 20, 28, and theexperimental signal 16 activates an electromagnetic radiation emittingdevice 22 such as an LED. The emissions 30 impinge on the target sample32 which in turn generates emissions 34 detected by the detection member36 such as a photodiode, all being similar to the prior embodimentillustrated in FIG. 1. The experimental output signal 38 passes througha preamplifier 40, and both the experimental output signal 38 and thereference output signal 26 pass through respective anti-aliasing infilters 42, 44 and analog-to-digital converters 48, 50. As in the priorembodiment, the output signals 38, 46, in digital form, are thencombined at the mixer 52. If the signals 38, 46 are not in quadrature,the DC component of the signal 54 is a number other than zero.

The AC component of signal 54 is removed by the low pass filter 56. Theoutput of the filter 56 is a DC error signal 58, the sign and magnitudeof which indicates the relative phase difference between signals 46 and38. As explained previously, the DC error signal 58 is preferably zerowhen signals 38 and 46 are in quadrature. The DC error signal 58 outputfrom the filter 56 is then directed back to the frequency calculator 64within the waveform generator 62 to simultaneously control both thefrequency and the phase of the output signals 16 and 26 as describedbelow.

In this embodiment of FIG. 2, the feedback error signal 58 causes thewaveform generator 62 to simultaneously change both the phase advance ofthe signal 26 relative to the signal 16 as well as the modulationfrequency of both signals 16 and 26. The phase and modulation frequencyare changed simultaneously until the DC error signal 58 indicates thatinput signals 46 and 38 are in quadrature, that is when the error signal58 is substantially zero. The phase and frequencies of the waveformsdetermined by the calculator 64 are indicated by binary numbers storedin the DSP 60. Although the waveform generator 62 and the containedfrequency and phase calculator 64 are digital, the high digitalresolution affords effectively continuous changes in frequency and phaseoffset. Thus, when the DC error signal 58 is substantially zero, thetime constant of the luminescence system can be calculated usingEquation 1. The phase delay through the sample 32 is simply 90°-N° withN° being the phase advance of the signal 26, and the frequency is knownfrom the digital number generated by the calculator 64.

The simultaneous and continuous variation of phase and frequency in afeedback loop acts to compress the phase and frequency ranges that arerequired for a particular luminescence lifetime range. Compared to theprior art techniques of Dukes and Venkatesh as disclosed above, thisFIG. 2 embodiment of the invention uses less expensive, more convenientcomponents that have narrower operating ranges. Moreover, while thisembodiment of FIG. 2 of the present invention employs continual andsimultaneous changes in phase offset and frequency of the output signals16 and 26, thereby sacrificing optimum lifetime measurement sensitivity,a heretofore unanticipated result is the benefit of frequency and phaserange compression for luminescence sensors.

This compression of the frequency and phase range over a wide lifetimerange is accomplished by using the continuously variable phase offsetand continuously variable frequency provided by the waveform generator62. Since the phase offset of the dual-output, variable-phase oscillator62 changes as the frequency changes, then a much smaller frequency rangeis needed. For example, if the phase-offset of the variable-phaseoscillator 62 were to change by 0.0038 degrees per Hz, then the entirerange between 1 μsec and 100 μsec can be covered with a frequency rangeof 19,900 Hz-3,900 Hz. This is a frequency range of only 5:1 as comparedto a range of 100:1 required by prior art devices and techniques usingvariable frequency and a predetermined, fixed phase offset. Thus, inthis particular embodiment illustrated in FIG. 2, both the phase andfrequency may vary and is known as "phase compression", for the use of acontinuously variable phase-offset compresses the frequency and phaseranges. This extends the measurable lifetime range for systems usinginexpensive, limited range components including but not limited tooscillators, waveform generators, amplifiers, and analog-to-digital anddigital-to-analog converters.

In the phase compression system 10 of FIG. 2, the frequency and phaseoutput of the multiple phase oscillator 62 is determined by a DC errorsignal 58 derived from the mixer 52. The error signal 58 from the mixer52 and low pass filter 56 controls the output frequency and phase of theoscillator 62 so that the two signals 38, 46 input to the mixer 52 areeventually in quadrature. This error signal 58 passes to the frequencyand phase calculator 64 which determines how the frequency and phaseshould change based on the error signal 58. The outputs of thecalculator 64 are binary numbers representing frequency and phase, andthese numbers are used by the waveform generator 62 to generate adigital representation of two sine waves, 16 and 26, at the frequencyand phase offset specified by the calculator 64.

While the waveform generator 62 creates a reference signal 26 that isadvanced with respect to the experimental signal 16 as with theembodiment of FIG. 1, the difference in this embodiment of FIG. 2 isthat the frequency of the signals 16 and 26 change continuously andsimultaneously with changes in the phase offset. In one preferredembodiment, the calculator 64 changes the frequency and phase accordingto the following relationship:

    N°=F·CF+N.sub.base                         Equation (2)

where N° is the phase offset between signals 16 and 26, F is thefrequency of signals 16 and 26, CF is the compression factor, andN_(base) is the base phase offset.

For example with CF=0.0038 deg/Hz, N_(base) =6.18 deg and a luminescencesensor lifetime of 101.6 μsec, the calculator 64 adjusts the systemfrequency to 4000 Hz, and the phase offset to 21°. These are theconditions where the error signal 58 is substantially zero. Withequivalent CF and N_(base) parameters, and a luminescent sensor lifetimeof 1.2 μsec, the calculator adjusts the system frequency to 19,900 Hz,and the phase offset to 82°. When the signals 38 and 46 are inquadrature, the lifetime of the sample 32 can be calculated using theknown phase and frequency. Since the action of the calculator 64compresses the frequency and phase range, inexpensive components withlimited range can be used in the present invention in place of expensiveand complex components.

In the above example, the frequency and phase range compression becausethe shortest lifetime to require a frequency of under 20,000 Hz.Currently, it is a distinct advantage to use DSP compatibleanalog-to-digital and digital-to-analog converters that have a maximumfrequency range of 20,000 Hz. This is due to the fact that such limitedfrequency range components are mass produced for consumer audioapplications and thus are inexpensive and simpler to use as compared towider frequency range components produced for more limited markets, e.g.scientific instrumentation.

A third embodiment of the present invention incorporates the concept ofdown conversion by mixing the reference and experimental signals withanother third signal of different phase and frequency, i.e. downconverting, to a fixed or variable lower frequency while preservingrelative phase information. In a simple lifetime measurement systemembodiment as illustrated in FIG. 1 and FIG. 2, the frequency at whichthe exciting light 30 is modulated and the frequency at which phasemeasurement takes place in the DSP 12 are essentially identical. Ashigher modulation frequencies are demanded by the measurement of shorterfluorescent or luminescent lifetimes, a point is reached where thenecessary program steps for phase comparison and correction cannot beexecuted between samples. This particular problem is overcome by theembodiments illustrated in FIGS. 3-4. In these embodiments, thehigh-frequency experimental and reference signals are each linearlymultiplied by a local oscillator frequency in a mixer. The resultingwaveform or signal is then filtered and presented to theanalog-to-digital converter.

Referring now with particularity to FIG. 3, the device 10 includes a DSP80 having a dual-output, variable-phase waveform generator 62. As in theprior embodiment, a frequency and phase calculator 64 determines theappropriate frequency and phase relationship of the two signals 16 and26 output by generator 62. The experimental signal 16 and the referencesignal 26 pass through their respective converters 20, 28. The signal 16activates an LED 22 which generates an emission 30 to impinge targetsample 32 to create an emission 34 which is detected by the photodiode36. The output signal 38 passes through the pre-amplifier 40.

In this particular embodiment, a second frequency generator 82 isdisposed within the DSP 80 and generates a signal 84 having a frequencydifferent from the frequencies of the output signals 38, 46. The signal84 passes through a digital-to-analog converter 86 and is then mixedwith the output reference signal 46 at a mixer 88 as well as with theoutput experimental signal 38 at yet another mixer 90. When the signal84 mixes with each of the signals 38, 46, a modified output referencesignal 94 and a modified output experimental signal 92 are created,respectively. Each of the signals 92 and 94 passes through theirrespective anti-aliasing filters 44 and 42, and analog-to-digitalconverters 50, 48 and are then demodulated at the digital mixer 52.

When the signal 84 is mixed with the reference signal 46, both the sumand the difference frequencies are incorporated into the modified outputsignal 92. Likewise, both the sum and difference frequencies of thesignal 84 and the signal 38 are reflected in the modified output signal94. The anti-aliasing filters 42 and 44 preferably remove the sumfrequency of the signals 84 and 46 and the sum frequency of the signals84 and 38, respectively, so that only the difference frequency of thesignals 84 and 46 and difference frequency of the signals 84 in 38 aremixed and compared at the mixer device 52. At the demodulator 52, thephases of the signals 92 and 94 are compared, and a feedback signal 54is generated by the mixer 52. This feedback signal 54 passes through thelow pass filter 56 and is then returned to the frequency and phasecalculator 64. The signal 58 is also directed towards the secondfrequency generator 82. As in the prior embodiment, the error signal 58indicates the sign and magnitude of the phase difference between signals92 and 94, and is preferably zero when these signals are in quadrature.The calculator 64 simultaneously changes the phase and frequency ofoutput signal 16 and 26, as in the prior embodiment of FIG. 2, such thata condition of quadrature is maintained at the mixer 52.

The difference frequencies of the modified output signals 92 and 94 areheld constant by action of the error signal 58 on the second frequencygenerator 82. The second frequency generator 82 tracks the signalfrequency output of the frequency generator 62 by always maintaining asignal frequency output that is different, i.e. higher or lower, by aconstant value, for example 10 kHz. Constant frequency inputs to thedemodulator 52 are preferred. One can anticipate a scheme, however,which sends variable frequency inputs to the demodulator 52 though thereis generally no benefit to such an implementation.

In evaluating this embodiment of FIG. 3, when a sinusoid of onefrequency linearly multiplies a sinusoid of another frequency, theresulting waveform or signal consists of a linear combination of a pairof sinusoids whose individual frequencies are the sum and difference ofthe two original frequencies. In a practical lifetime measurementcircuit as in the embodiment of FIG. 3, the sum frequency is rejected bya filter, and the difference frequency which may be quite low is passedto the analog-to-digital converters for further processing within theDSP 80 as described above. The difference frequency can have anyconvenient value, and it is determined only by the relationship betweenthe signals 38 and 46 frequency and the signal 84 frequency. The phaserelationship between the high frequency reference signal 46 and theexperimental signal 38 are maintained through the down conversionprocess.

Another embodiment of FIG. 3 uses external digital waveform generatorsin place of the component generators 62 and 82 and the D/A converters20, 28 and 86. This particular embodiment would be used when thefrequencies of the signals 16 and 26 are too high to be generatedinternally within digital signal processor 80. In this case the externalgenerators would preferably consist of single chip waveform generatorswhich would be controlled by the digital signal processor 80 and derivetheir clock frequency from the same analog oscillator as digital signalprocessor 80.

Referring now to FIG. 4, this embodiment imposes an additionalrequirement and capability on the down conversion process as compared tothat of FIG. 3 explained above. As a part of measuring the phasedifference between the experimental and reference signal sinusoids, theDSP of this FIG. 4 executes a program that implements an additionalnumerical direct digital synthesis frequency generator, the numericaldirect digital synthesis generator being used in all the priorembodiments as the devices 14 and 64. Referring to FIG. 4, the downconversion arrangement of FIG. 3 remains substantially the same.However, the experimental signal 16 and the reference signal 26 aregenerated by one single-output frequency generator 96 and are generatedat the same phase and frequency.

In one embodiment as illustrated in FIG. 4, the frequency generators 96and 82 are specialized DSP components that are external to the maindigital signal processor 98 and contain digital-to-analog converters 142and 140, respectively. In another embodiment, the frequency generators96 and 82 may be internal to the main DSP 98, as shown for example inFIGS. 1 and 2. While it is actually preferred that the frequencygenerator 96 is implemented internally within the DSP 98, one would thenpreferably then use the downconversion single quad technique of FIG. 2.When the signals 92 and 94 in this embodiment of FIG. 4 pass through theanalog-to-digital converters 50 and 48, they are not mixed directlytogether as with the prior embodiments. Instead, an additional dualoutput, multiphase digital synthesis frequency generator 100 is providedwithin the DSP 98.

In preferred form, the dual-output, multi-phase digital synthesisfrequency generator 100 includes a frequency and phase calculator 102that generates a first internal signal 104 and a second internal signal106, each of which has a frequency which matches exactly the frequencyof the signals 92 and 94, which is the difference frequency between thefrequency generated by the generator 96 and the frequency generated bythe generator 82. The first internal signal 104 is generated such thatit has a phase relative to input signal 92 of 90°. This is accomplishedby mixing the signals 92 and 104 at an internal mixer 108. The output ofthe mixer 108 is directed to a low pass filter 150 which outputs anerror signal 152, the sign and magnitude of which indicates the relativephase difference between signals 104 and 92. The error signal 152,preferably zero when the signals 92 and 104 are in quadrature, isdirected to the frequency and phase calculator 102. The calculator 102then adjusts the phase of the signal 104 until the error signal 152indicates that signals 92 and 104 are in quadrature.

At the same time, the signal 94 is directed toward another internalmixer 110, and the frequency generator 100 generates the second internalsignal 106 of preferably identical frequency with the signals 92, 94 and104, and with a phase that is advanced a known amount with respect tothe phase of the signal 104. The signal 106 is mixed with the signal 94at the internal mixer 110, the output of which is directed through a lowpass filter 56 to create another internal error signal 114 which isdirected to the frequency and phase calculator 102. Based on the signand magnitude of the error signal 114, the phase of the signal 106 isshifted until the signal 106 and the signal 94 are in quadrature at themixer 110. Since the signals 92 and 94 preferably differ in phase onlybased on the phase shift caused by the target sample 32, each of thesignals 92 and 94 are individually placed into quadrature with separatesignals in order to determine this difference in phase at the synthesisfrequency generator 100. The phase difference between the signals 104and 106 thus reflects the phase difference between the input signals 92and 94. The fluorescence lifetime of the sample can be calculated usingthe measured phase shift and frequency with Equation 1.

In this embodiment of FIG. 4, the frequency generators 82 and 96 areinexpensive, small single integrated circuit, commercially availablecomponents. These external generators 82 and 96 do not, however, providea means for communicating the current phase of the output signals 16 and26 to the DSP 98. As a result, the digitized reference signal 92 is atsome unknown phase. The additional internal phase lock loop, which ismade up of the mixer 108, the filter 150, the frequency generator 100and the frequency calculator 102, generates a signal 104 that is phaselocked to the input signal 92. The signal 104 then becomes the phasereference for the second mixing process using mixer 110.

The following example illustrates the purpose and function of theadditional phase locked loop of FIG. 4. The signal 92 is digitized at 50and has some unknown phase which we designate "α". Simultaneously, thesignal 94 is digitized at 48 and differs in phase from the signal 92 by"β", the result of the phase shift caused by the sensor 32 and any phaseshifts due to the analog components 22, 36 and 40. Thus, the signal 94has a phase of α+β as indicated in FIG. 4. The phase locked loop whichincludes the mixer 108 generates the signal 104 with a phase shift ofα+90°. The frequency generator 100 then creates a signal 106 that has anadded phase shift of "δ" relative to the signal 104. Therefore, thesignal 106 has a phase of α+90+δ. The mixer 110 and error signal 114impose on the signal 106 the condition that it must be in quadraturewith the signal 94. This is accomplished by adjusting δ, the amount ofadditional phase shift relative to signal 104. At the mixer 110 we findthat the signals 94 and 106 differ by 90°, that is the phase of thesignal 94 plus 90°, α+β+90, is equal to the phase of signal 106 which weknow to be α+90+δ, or

    α+β+90=α+90+δ.

Simplifying the above, we find that β=δ. The amount of known phase shiftadded to the signal 106, δ, is equal to the phase shift caused by thefluorescence experiment 32, along with the other analog components 22,36, and 40. Finally the phase shift indicated by δ may be used withEquation 1 to calculate the fluorescence lifetime.

An alternative application of the embodiment of FIG. 4 includes afrequency feedback signal 154 which passes from the frequency and phasecalculator 102 for varying the output of the frequency generators 82 and96. In this manner, the frequencies of the signals 16, 26 and 84, andthe phase of the signal 106 may also be simultaneously varied as in theembodiment illustrated in FIG. 2. Another alternative embodiment of FIG.4 includes a digital signal processor 98 which contains an internalfrequency generator 96 and digital to analog converter 140.

Referring now to FIG. 5, this embodiment is similar to that shown inFIG. 4 except that it lacks a means for downconversion of high frequencysignal to lower frequencies for quadrature phase detection. Referring toFIG. 5, the down conversion arrangement of FIG. 4 has been eliminated.This FIG. 5 embodiment is particularly useful when only a single phasedigital waveform generator is available in place of a dual-phase outputdigital waveform generator. In one implementation of this embodiment asillustrated in FIG. 5, the frequency generator 96 is a specialized DSPcomponent that is external to the main digital signal processor 98 andcontains a digital to analog converter 142. In another implementation,the frequency generator 96 is internal to the main DSP 98, as shown forexample in FIGS. 1 and 2.

In this embodiment, as in the prior embodiment, the experimental signal16 and the reference signal 26 are generated by one single-outputfrequency generator 96 and are generated at substantially the same phaseand frequency. As in prior embodiments, the experimental signal 16passes through the light source 22, the luminescent sample 32, and thephotodetector 36. The signal 38 output from the photodetector 36 isconverted to a voltage at the preamplifier 40 and filtered at theanti-aliasing filter 42 as in prior embodiments. The experimental outputsignal 38 is then digitized at the analog-to-digital converter 48. Thereference input signal 26 also passes through a substantially identicalanti-aliasing filter 44 and is digitized at the analog-to-digitalconverter 50. The digitized representations of the experimental outputsignal 38 and the reference signal 26 are digital experimental signal 94and digital reference signal 92, respectively.

As in the prior embodiment of FIG. 4, when the signals 92 and 94 passthrough the analog-to-digital converters 50, and 48, they are not mixeddirectly together. Instead, an additional dual-output, multiphasedigital synthesis frequency generator 100 is provided within the DSP 98.As in the prior FIG. 4 embodiment, this generator 100 allows both thedigital experimental signal 94 and the digital reference signal 92 to becompared in quadrature at two different digital mixers, 108 and 110.

As in the previously described embodiment of FIG. 4 and in preferredform, the dual-output, multi-phase digital synthesis frequency generator100 includes a frequency and phase calculator 102 that generates a firstinternal signal 104 and a second internal signal 106, each of which hasa frequency which matches substantially exactly the frequency of thesignals 92 and 94, which is the difference frequency between thefrequency generated by the generator 96 and the frequency generated bythe generator 82. The signal 104 is generated such that it has a phaseof 90° relative to the input signal 92. This is accomplished by mixingthe signals 92 and 104 at a mixer 108. The output of the mixer 108 isdirected to a low pass filter 150 which outputs error signal 152, thesign and magnitude of which indicates the relative phase differencebetween signals 104 and 92. The error signal 152, preferably zero whenthe signals 92 and 104 are in quadrature, is directed to the frequencyand phase calculator 102. The calculator 102 adjusts the phase of signal104 until the error signal 152 indicates that signals 92 and 104 are inquadrature.

At the same time, the signal 94 is directed toward a mixer 110, andgenerator 100 generates signal 106 of preferably identical frequencywith signals 92, 94 and 104, and with phase that is advanced a knownamount with respect to the phase of signal 104. The signal 106 is mixedwith the signal 94 at the mixer 110, the output of which is directedthrough the low pass filter 56 to create the error signal 114 which isdirected to the frequency and phase calculator 102. Based on the signand magnitude of the error signal 114, the phase of the signal 106 isshifted until the signal 106 and the signal 94 are in quadrature at themixer 110. Since the signals 92 and 94 preferably differ in phase onlybased on the phase shift caused by the target sample 32, each of thesignals 92 and 94 are individually placed into quadrature with separatesignals in order to determine this difference in phase at the synthesisfrequency generator 100. The phase difference between signals 104 and106 thus reflects the phase difference between the input signals 92 and94. The fluorescence lifetime of the sample can be calculated using themeasured phase shift, frequency with Equation 1.

As with the prior embodiment, the additional phase locked loop made upof the generator 100, the feedback signal 152, the mixer 108 and theintegrator 150 allows an additional known amount phase shift to be addedto internal reference signal 106 so that the digitized experimental canbe compared in quadrature to a signal of known phase.

An alternative application of the embodiment of FIG. 5 includes thefrequency feedback signal 154 which passes from the frequency and phasecalculator 102 for varying the output frequency of the generator 96. Inthis manner, the frequencies of the signals 16 and 26, and the phase ofsignal 106 may also be simultaneously varied as in the embodimentillustrated in FIG. 2.

Referring now to FIG. 6, this embodiment describes a device 12 that usesa Digital Signal Processor 200 for measuring phase shifts of an analogsignal 202 through a phase shifting element 204, relative to the phaseshifts of an analog signal 206 of substantially identical frequencythrough a reference element 208. In a preferred embodiment, the analogsignals 202 and 206 are sinusoidal and substantially identical infrequency. The signals 202 and 206 are preferably generated by adual-output, variable phase and frequency waveform generator 210, whichis contained in the DSP 200. The digital outputs, 212 and 214, of thewaveform generator 210 are directed to digital-to-analog converters 216and 218. The digital-to-analog converters output analog signals 202 and206 which are directed to the phase shifting element 204 and thereference element 208, respectively.

In the preferred embodiment of this FIG. 6, the phase shifting element204 contains a fluorescent material that changes lifetime in response tosome analyte. Additionally, the phase shifting element 204 may alsocontain an excitation light source, a photodetector, a pre-amplifier,and anti-aliasing filters as described in the previous embodiments. Thereference element 208 preferably contains substantially identicalanti-aliasing filters which add substantially the same amount of phaseshift to the signal as the anti-aliasing filters of the phase shiftingelement, as explained in the previous embodiments. It should be notedthat the phase shifting element does not necessarily contain afluorescent sample. It may, in fact, consist of many types of electricalor optical phase shifting components.

The output signal 220 of the phase shifting element 204, and the outputsignal 222 of the reference element 208 are directed then towardsanalog-to-digital converters 224 and 226, respectively. The analog todigital converters 224 and 226 convert the analog signals 220 and 222into digital representations in the DSP 200. The digitized signals 220and 222 are then directed towards a digital phase demodulator 228. Inthe preferred embodiment of this FIG. 6, the DSP 200 contains a digitalphase demodulator 228, a low pass filter 230, a dual-outputvariable-phase waveform generator 210, and a filtered feedback errorsignal 232 from the phase demodulator 228. As described in previousembodiments, these elements act in concert to force the digital signals220 and 222 into quadrature at the digital phase demodulator 228. Underconditions of quadrature, the phase shift between the two signals 220and 222 due to the phase shifting element 204 can be determined usingone of the methods of the previous embodiments.

This FIG. 6 embodiment of the invention preferably includes a singleanalog timing element 234, which provides a master timing base for alldigital signal generation and phase comparison operations in the DSP200. In the preferred embodiment, the timing element 234 consists of aquartz crystal oscillator. The timing element 234 generates a highfrequency clock signal 236 that is directed to a clock divider 238. Inone preferred embodiment, the high frequency clock signal 236 isapproximately 25 MHz. The clock divider 238 then digitally divides theclock signal 236 into a lower frequency clock signal 240. This lowerfrequency clock signal 240 becomes the timing signal for all operationsrelating to the determination of the relative phase between signals 220and 222. In the preferred embodiment, the frequency of the clock signal240 is substantially 48 kHz. Depending on the specific components used,the clock signal 240 may differ significantly from 48 kHz.

The clock signal 240 is preferably directed to the digital-to-analogconverters 216 and 218 and the analog-to-digital converters 224 and 226.At the digital-to-analog converters 216 and 218, the clock signal 240causes the conversion of a pair of digital points representing waveforms212 and 214 to analog signals 202 and 206. Simultaneously, the clocksignal 240 causes the analog-to-digital converters 224 and 226 toconvert the incoming analog signals 220 and 222 into a pair of digitalpoints representing the analog signals 220 and 222. Since the analogsignal generation and digitization are synchronized to the clock signal240, these events occur simultaneously and with essentially no phasejitter.

The clock signal 240 additionally causes the phase demodulator 228, thefilter 230 and the waveform generator 210 to perform calculations on thenext set of digital numbers 220 and 222. When the analog signals 220 and222 are digitized, their digital representations are first directed to adigital phase demodulator 228. The result of the digital phasedemodulator 228 is directed to a digital filter 230, which outputs afiltered error signal 232 which is then directed to the waveformgenerator 210. The waveform generator 210 then generates a new set ofdigital numbers for digital signals 212 and 214. The phase and frequencyof the digital signals 212 and 214 are determined by the value of theerror signal 232. At every cycle of the clock signal 240, the aboveoperations are performed once and the operations are completed beforethe next cycle of clock signal 240. The waveform generator 210 alsoprovides for a means that the frequency and relative phase of signals212 and 214 can be output at each cycle of clock signal 240. Thus, ateach cycle of the clock signal 240, the lifetime of the phase shiftingelement 204 can be determined essentially continuously using Equation 1.

In the preferred embodiment of FIG. 6, the clock divider 238, thedigital-to-analog converters 216 and 218 and the analog-to-digitalconverters 224 and 226 are contained in a integrated circuit separatefrom the DSP 200, while the waveform generator 210 is contained withinthe integrated circuit of the DSP. It should be understood that thesecomponents may be contained either within or outside of the DSP 200without departing from the spirit of the invention. Moreover, thefrequencies of the signals 212 and 214 are substantially the same as thefrequencies of signals 220 and 222. It should also be further understoodthat the phase shifting element 204 and the reference element 208 caninclude a means for downconverting, as previously described, from theinput frequencies 202 and 206 to lower frequencies for signals 220 and222 without departing from the spirit of the invention.

In order to better understand how various of the embodiments of thepresent invention operate, the following examples are provided. Itshould be understood, however, that these examples are only for purposesof illustration and are not intended to limit the scope of the inventionwhich is defined by the claims appended hereto.

EXAMPLE I

The device 10 of FIG. 1 was implemented using a commercially availableADSP2181 EZLAB prototyping kit from Analog devices and additional analogcomponents as described below. The DSP 12 consisted of an Analog DevicesADSP-2181 KS-133. Dual output waveform generator 14 was implemented insoftware using Direct Digital Synthesis, a commonly used method forgenerating digital waveforms (see description in Analog Devicestechnical specifications for part#AD9830, Rev. A, p. 10). The twodigital output signals 16 and 26 were directed to a ΔΣ Stereo (2channel) CODEC (Analog Devices part #AD1847JP) which generated two 20KHz counterpart analog sine-waves with a relative phase difference asspecified by the waveform generator 14. The CODEC output each analogsignal with an sampling rate of 48 KHz.

One 20 KHz sine wave 16 was directed to an operational amplifier (AnalogDevices AD810) that provided sinusoidal current drive to the LED 22. Thelight output of the blue LED (Nichia NSPB500S) was immediately filteredusing a blue-interference filter that blocked the longer wavelengthlight (yellow, orange and red) produced by the LED. The resulting bluelight, 30, was directed towards a sample 32.

The sample 32 used in this example and embodiment consisted ofplatinum-tetrapentafluorophenyl porphyrin (PtTFPP) dispersed in aproprietary oxygen permeable matrix. This sample had a luminescentlifetime of 18.5 microseconds at ambient temperature and pressure. Thered luminescence of the sample 34 was directed towards a photodiode 36having a red interference filter to remove any scattered blue excitationlight 30. The output current 38 of the photodiode 36 (Hamamatsu PINphotodiode S4707-01), was directed towards a transimpedance amplifier 40(Burr Brown OPA655) with gain which converted the sinusoidally varyingphotodiode output current into a sinusoidally varying voltage. Thevoltage signal 38 was directed to an anti-aliasing filter 42. Theanti-aliasing filters 44 and 42 consisted of single-section, low-pass RCfilters with time constants of 3.3 μsec. The output of the anti aliasingfilter 42 was directed towards the input side 48 of the Stereo CODECwhere the signal was sampled and digitized at a rate of 48 KHz.

The analog reference signal 26 was directed to an anti-aliasing filter44, which consisted of essentially the same components as theanti-aliasing filter 42 in the signal path as described above. Thefiltered reference signal 46 was directed to the second input 50 of theCODEC and digitized at a rate of 48 KHz, synchronously with the sampleanalog signal 38.

The digitized representations of the signals 46 and 38 were multipliedpoint by point at a rate of 48 KHz at phase demodulator 52. The phasedemodulator 52, implemented in software, multiplied the digitized datapairs of the time series generated by the CODEC 48, 50. The result ofthe phase demodulator 52 was sent to a low pass filter 56. The low passfilter 56 consisted of a digital IIR single or double pole low passfilter implemented in the ADSP2181 (see Oppenheim, A. V., and R. W.Schafer. Discrete-Time Signal Processing. Englewood Cliffs, N.J.:Prentice-Hall, 1989.)

The output 58 of this filter 56, which represents the sign and magnitudeof the phase difference between the signals 46 and 38, was directed tothe dual output waveform generator 14. The error signal 58 causes thewaveform generator 14 to change the phase advance of the referencesignal 26 to a value which puts the signals 46 and 38 in quadrature atthe phase demodulator 52. This condition is met when the error signal iszero.

Using the sample described above at ambient temperature and pressure,and 20 KHz excitation, the sample produced a phase shift of -21.9°.Several other elements contribute phase shift equally to both the samplechannel and the reference channel, (e.g. the CODEC and the anti-aliasingfilters) and did not change the relative phase of the two signals. Sincethere was a -21.9° phase shift though the luminescent sample, the dualoutput waveform generator 14 adjusted the phase advance of referencesignal 26 to 68.1° to achieve quadrature conditions at the phasedemodulator 52. The phase shift of the sample was determined bycomputing the difference between 90° (quadrature) and the phase advanceadded to the reference channel, 68.1°, or in other words90°-68.1°=21.9°. The lifetime of the sample 32 was calculated usingEquation 1 above.

EXAMPLE II

The device 10 of FIG. 2 was implemented using a commercially availableADSP2181 EZLAB prototyping kit from Analog devices and additional analogcomponents as described below. The DSP 60 consisted of an Analog DevicesADSP-2181 KS-133. Dual output waveform generator 14 was implemented insoftware using Direct Digital Synthesis as in Example I. The two digitaloutput signals 16 and 26 were directed to a ΔΣ Stereo (2 channel) CODEC(Analog Devices part #AD1847JP) which produced two counterpart analogsine-waves with a relative phase difference and frequency as specifiedby the waveform generator 14. For any frequency and phase relationship,the CODEC output the analog signal using a sampling rate of 48 KHz.

One sine wave 16 was directed to an operational amplifier (AnalogDevices AD810) that provided sinusoidal current drive to the LED 22. Thelight output of the blue LED (Nichia NSPB500S) was immediately filteredusing a blue-interference filter that blocked the longer wavelengthlight (yellow, orange and red) produced by the LED. The resulting bluelight 30 was directed towards a sample 32. The sample 32 used in thisExample II was similar to that of Example I and consisted ofplatinum-tetrapentafluorophenyl porphyrin (PtTFPP) dispersed in aproprietary oxygen permeable matrix. However, this sample had aluminescent lifetime of 7.27 microseconds at 100% oxygen at ˜760 Torrand 45° C.

The red luminescence 34 of the sample 32 was directed towards aphotodiode 36 (Hamamatsu PIN photodiode S4707-01) having a redinterference filter to remove any scattered blue excitation light 30.The output current of the photodiode 36 was directed towards atransimpedance amplifier 40 (Burr Brown OPA655) with gain whichconverted the sinusoidally varying photodiode output current into asinusoidally varying voltage. The voltage signal 38 was directed to ananti aliasing filter 42. The anti-aliasing filters 44 and 42 consistedof single-section, low-pass RC filters with time constants of 3.3 μsec.The output of the anti-aliasing filter 42 was directed towards the inputside 48 of the Stereo CODEC where the signal was sampled and digitizedat a rate of 48 KHz.

The analog reference signal 26 was directed to an anti-aliasing filter44 which consisted of essentially the same components as theanti-aliasing filter 42 in the signal 38 path. The filtered referencesignal 46 was directed to the second input 50 of the CODEC and digitizedat a rate of 48 KHz synchronously with the sample analog signal 38.

The digitized representations of the signals 46 and 38 were multipliedpoint by point at a rate of 48 KHz at the phase demodulator 52. Thephase demodulator 52, implemented in software, multiplied the digitizeddata pairs of the time series generated by the CODEC. The resultingsignal 54 of the phase demodulator 52 was sent to a low pass filter 56.The low pass filter consisted of a digital IIR single or double pole lowpass filter implemented in the ADSP2181 as previously mentioned forExample I. The output signal 58 of this filter 56, which represents thesign and magnitude of the phase difference between signals 46 and 38,was directed to the dual output waveform generator 14. The error signal58 caused the waveform generator 14 to simultaneously and continuouslychange the phase advance of the reference signal 26 and the frequency ofsignals 16 and 26 to values which puts the signals 46 and 38 inquadrature at the phase demodulator 52. This condition is met when theerror signal is zero.

Using the sample 32 described above in 100% oxygen at 760 Torr and 45°C., a 15.940 kHz excitation and phase offset of 53.904° were required toachieve conditions of quadrature at the mixer 52. Several other elementscontributed phase shift equally to both the sample channel and thereference channel, (e.g. the CODEC, and the anti-aliasing filters) andthus did not change the relative phase of the two signals. As previouslydescribed, the dual output waveform generator 14 contains a frequencycalculator and phase calculator 64 which determines the phase advance ofsignal 26 and the frequency of signals 16 and 26 based on the feedbackerror signal 58. The frequency and phase calculator were set to adjustthe phase and frequency according to the following equation:

    N°=0.0038 F+6.18                                    Equation (3)

The phase advance of the signal 26, that is N°, and the frequency F ofthe signals 16 and 26 at any particular phase advance, N°, weredetermined by Equation 3. The waveform generator 62 increased ordecreased N°, and hence simultaneously lowered or raised the frequency Faccording to Equation 3, until the error signal indicated that theinputs 38 and 46 to the phase demodulator 52 were in quadrature.

Since there is a -36.096° phase shift though the luminescent sample, thedual output waveform generator adjusted the phase advance of referencesignal 26 to 53.904° to achieve quadrature conditions at the phasedemodulator 52. Following Equation 3, the frequency was set to 15.94kHz. The phase shift of the sample was determined by computing thedifference between the phase advance added to the reference channel,53.904°, and 90° (quadrature), or 53.904°-90°=-36.096°. The lifetime ofthe sample then was calculated using Equation 1 above.

EXAMPLE III

The device 10 of FIG. 3 is implemented using a commercially availableADSP2181 EZLAB prototyping kit from Analog devices and additional analogcomponents as described below. The DSP 60 consists of an Analog DevicesADSP-2181 KS-133. Dual output waveform generator 62 is implemented insoftware using Direct Digital Synthesis as previously mentioned in theExamples I and II, or if the output frequencies required are too highfor generation in software, the waveform generator is implementedexternally in a custom DSP chip and clocked by the same master analogclock as the other digital components. The waveform generator 62generates two digital output signals 16 and 26 that are then directed toan appropriate digital-to-analog converter 20 which produces twocounterpart analog sine-waves with a relative phase difference andfrequency as specified by the waveform generator 62.

As in the previous examples, one sine wave 16 is directed to anoperational amplifier (Analog Devices AD810) to provide sinusoidalcurrent drive to the LED 22. The light output of a blue LED (NichiaNSPB500S) is immediately filtered using a blue-interference filter thatblocks the longer wavelength light (yellow, orange and red) produced bythe LED. The resulting blue light 30 is then directed towards a sample32. The sample 32 consists of a fluorescent sample the lifetime of whichis quenched in the presence of the analyte of interest. The lifetime ofthe sample can be quite short, for example 0.5 nsec to 5 nsec. At 10 MHzexcitation, a fluorescent sample with a 5 nsec lifetime exhibits a phaseshift of 17°.

The longer wavelength fluorescence of the sample 34 is directed towardsa photodetector 36, such as a photomultiplier tube, avalanche photodiodeor other appropriate detector, having an appropriate interference filterto remove any scattered blue excitation light 30. The output current 38of the photodetector 36 is directed towards a transimpedance amplifier40 (e.g. Burr Brown OPA655) to convert the sinusoidally varyingphotodetector output current into a sinusoidally varying voltage 38. Thevoltage signal 38 is then directed to an analog mixer 90. The analogreference signal 26 is also directed to similar analog mixer 88.

This embodiment provides for a second waveform generator 82, which inthis example is a commercially available AD9830. This waveform generator82 outputs a high frequency digital signal 84 which is then convertedinto an analog signal at an appropriate digital-to-analog converter 86as previously described. The digital-to-analog converter 86 may in factbe integral to the second waveform generator 82. The second waveformgenerator 82 outputs a signal 84 with a frequency that is a constantdifference from the frequencies of signals 16 and 26. For example, ifthe signals 16 and 26 are 10 MHz signals, the second waveform generator82 outputs a signal of 10.02 MHz. In this example, the constantdifference between the first generator 62 and second generator 82 is 20KHz. If the output frequency of the waveform generator 62 changes, theoutput of the second generator 82 tracks the output frequency of thefirst generator 62 such that the difference in frequencies remains 20kHz.

The analog output 84 of the second generator is split into two identicalsignals directed towards the analog mixers 88 and 90. Both analog mixers88 and 90 perform substantially identically, their outputs being alinear multiplication of the two input signals. These outputs 92 and 94contain the sum and difference frequencies of the input signals to therespective mixers 88 and 90. In this example, the mixer output thusconsists of a 20.02 MHz signal and a 20 kHz signal.

The sum and difference frequency outputs of the analog mixers 88 and 90are filtered at similar anti-aliasing filters 44 and 42. Theseanti-aliasing filters 44, 42 are configured so that the sum frequencies,20.02 MHz, are removed, leaving only the difference frequencies, i.e. 20kHz. As described above, the downconverted 20 kHz difference frequenciescarry the same relative phase information as did the 10 MHz signals thatpassed through the fluorescent sample and the reference path.Appropriate anti-aliasing filters are, for example, single-section,low-pass RC filters with time constants of 3.3 μsec.

The output signal of each of the anti-aliasing filters 42 and 44 isdirected towards the input of an analog-to-digital converter 48, 50,respectively, where the signal is sampled and digitized at anappropriate rate, e.g. 48 KHz. The digitized representations of filteredsignals 92 and 94 are then multiplied point by point at a rate of 48 KHzat the phase demodulator 52. The phase demodulator 52, implemented insoftware, multiplies the digitized data pairs of the time seriesgenerated by the analog-to-digital converter. The resulting signal 54 ofthe phase demodulator 52 is sent to a low pass filter 56. The low passfilter consists of a digital IIR single or double pole low pass filterimplemented in the ADSP2181 as previously mentioned.

The output signal 58 of this filter 56, which represents the sign andmagnitude of the phase difference between filtered signals 92 and 94, isdirected to the high-frequency dual-output waveform generator 62 as wellas the second high frequency generator 82. The error signal 58 causesthe waveform generator 62 to simultaneously and continuously change thephase advance of the reference signal 26 and the frequency of signals 16and 26 to values which puts the filtered signals 92 and 94 in quadratureat the phase demodulator 52. This condition is met when the error signalis zero.

The dual output waveform generator 62 can be operated in one of twomodes. It can be operated in a first mode at a constant frequency andvariable phase shift, as in Example I, or it can be operated in a secondmode simultaneously at a continuously variable frequency and phase withcompression, as in Example II. If the waveform generator 62 is operatedin the second mode, the error signal 58 also acts on the second highfrequency generator 82 to adjust its frequency so that it is at aconstant 20 kHz difference from the frequency output of the firstgenerator 62. The sample lifetime is calculated as in Examples I and IIusing the phase shift and frequency with Equation 1.

EXAMPLE IV

The device 10 of FIG. 4 was implemented using a commercially availableADSP2181 EZLAB prototyping kit from Analog devices and additional analogcomponents as described below. The DSP 98 consisted of an Analog DevicesADSP-2181 KS-133. Unless stated otherwise, the components of thisExample IV were the same specific components utilized in the previousexamples. Instead of the dual-output waveform generator described in theExample III above, a single-output Direct Digital Synthesis waveformgenerator 96 was used to generate a high frequency sinusoid that drovean LED 22 to generate blue light for excitation of the sample. Thewaveform generator 98 consisted of an Analog Devices EVAL-AD9830EB whichcontained an analog Devices AS9830 Direct Digital Synthesis IC.

The waveform generator 96 contained an internal analog-to-digitalconverter 142 and output a sine wave of a known frequency, i.e. 1.013MHz. This sine wave output was split into two identical signals 16 and26. One signal 16 was directed towards an operational amplifier, AnalogDevices AD811, which provided sinusoidal current drive to a blue LED 22(Nichia NSPB500S), and the other signal, the analog reference signal 26,was sent to an analog mixer 88.

The light output of a blue LED 22 (Nichia NSPB500S) was immediatelyfiltered using a blue-interference filter that blocked the longerwavelength light (yellow, orange and red) produced by the LED. Theresulting blue light 30 was directed towards a sample 32. The sample 32consisted of a dilute fluorescent sample in a buffer of pH 7.6. Thelifetime of the sample was 4 nsec under ambient conditions.

The longer wavelength luminescence of the sample 34 was directed towardsa photomultiplier tube 36 (Hamamatsu R5600U-01) which had a 600 nmlongpass filter to remove any scattered blue excitation light 30. Theoutput current 38 of the photomultiplier tube 36 was directed towards atransimpedance amplifier 40 (e.g. Burr Brown OPA655) to convert thesinusoidally varying photodetector output current into a sinusoidallyvarying voltage. The voltage signal 38 was directed to an analog mixer90. The analog reference signal 26 was directed to a similar analogmixer 88.

This device 10 had a second waveform generator 82, which consisted of aAnalog Devices EVAL-AD9830EB containing an analog Devices AS9830 DirectDigital Synthesis IC. This second waveform generator 82 output a highfrequency analog signal 84 that was a constant difference from thefrequencies of signals 16 and 26. In this example the second oscillatorfrequency was 1.0078 MHz, a constant difference from the firstoscillator frequency of 5.2 kHz. The analog output 84 of the secondgenerator 82 was split into two identical signals directed towards theanalog mixers 88 and 90. Both analog mixers 88, 90 acted substantiallyidentically in that the output of each was a linear multiplication oftheir two respective input signals, i.e. signals 26 and 84 for the mixer88 and signals 38 and 84 for the mixer 90. These outputs 92 and 94contained the sum and difference frequencies of the input signals. Inthis example, the mixer output consisted of a 2.0208 MHz signal and a5.2 kHz signal.

The sum and difference frequency outputs of the analog mixers 88, 90were filtered at similar anti-aliasing filters 44 and 42. Theseanti-aliasing filters were configured so that the sum frequencies,2.0208 MHz, were removed, and the difference frequencies, i.e. 5.2 kHz,were passed on. As described above, the downconverted 5.2 kHz differencefrequencies carried the same relative phase information as did the 1.013MHz signals 16, 26 that passed through the luminescent sample 32 andthat acted as the analog reference signal, respectively. Single-section,low-pass RC filters with time constants of 3.3 μsec were used to filterthe downconverted signals.

The output of the anti-aliasing filters 44 and 42 were directed to theinputs 50, 48 of a ΔΣ Stereo (2 channel) CODEC (Analog Devices part#AD1847JP) where the signals were digitized at a rate of 48 KHz. Thedigitized filtered reference signal 92 was directed to a digital phasedemodulator 108. A second input to the digital phase demodulator was a5.2 kHz digital sine wave 104, generated by an internal dual-outputwaveform generator 100. The digital phase demodulator 108 consisted of apoint by point multiplication which operated at 48 kHz, which is therate of digitization of the CODEC.

The error signal generated by this phase demodulator 108 was thenfiltered with a digital IIR single or double pole low pass filter 150implemented in the ADSP2181 as previously mentioned. The filtered errorsignal 152 was input to the dual-output waveform generator 100, whichadjusted the phase of the digital sine wave 104 such that it was inquadrature with the digitized reference signal 92 at the demodulator108. Because the reference signal 92 has an unknown phase of α, as notedin FIG. 4, the digital phase lock loop described above locks one output,104, of the internal dual-output waveform generator to the digitizedreference input, signal 92. Specifically, the action of the mixer 108and the feedback error signal 152 causes signal 104 to have a phaseshift 90° advanced with respect to the signal 92, or α+90°.

The 5.2 kHz downconverted digitized signal from the fluorescenceexperiment, signal 94, was directed towards a second digital phasedemodulator 110 in the DSP. The second input to this phase demodulatorwas the second output 106 of the internal dual-output waveform generator100. As with the other digital phase demodulator 108, the result of thepoint-by-point multiplication passes through a IIR single or double polelow pass digital filter 56. This filtered signal 114, the error signalof the phase demodulator 110, was directed towards the phase andfrequency calculator 102 of the internal dual-output waveform generator100. The phase and frequency calculator 102 caused the waveformgenerator 100 to output a signal 106 at 5.2 kHz with a phase 90°advanced with respect to the digitized signal from the fluorescenceexperiment, signal 94.

Because both signals 104 and 106 generated by the internal dual-outputwaveform generator 100 are in quadrature with the downconverteddigitized reference and fluorescence experiment signals 92 and 94,respectively, the phase difference between the two signals generated bythe internal dual-output waveform generator 100 reflect the phasedifference between the digitized reference and fluorescence experimentsignals 92, 94. Thus, the total phase delay resulting from thecontribution of the excitation light source, the fluorescenceexperiment, the photodetector and the transimpedance amplifier areknown. Since the phase delay added by the fluorescence experiment 32 isthe parameter of interest, the phase delays due to other components ofthe system were factored out by measuring a fluorescence sample of knownlifetime.

In this particular example, the phase delays of the excitation lightsource, fluorescent sample (fluorescein), the photodetector and thepre-amplifier amounted to -13.42°. This was measured by determining thedifference between the two digital output signals 104, 106 of theinternal dual-output waveform generator 100, while both of the phasedemodulators 108, 110 in the DSP were operating in quadrature. Of the-13.42° phase shift, -1.46° were due to the 4 ns lifetime of thefluorescent sample, and -11.96° resulted from phase shift in theexcitation, photodetector and trans-impedance amplifier. Thenon-fluorescence sample phase shifts of -11.96° were assumed to beconstant and thus subtracted from the measured phase difference betweenthe downconverted signals 92 and 94. Equation 1 was used with θ=1.46°,and f=1.013 MHz to calculate the fluorescence lifetime of the sample.

As can be seen from the above, it is clear that the present inventionprovides a simple and effective apparatus and method for measuringenvironmentally and medically important analytes. The present inventionaccomplishes this by providing a unique apparatus and method formeasuring time delay in samples targeted by electromagnetic radiationand in particular fluorescence emissions. The present invention providesa fluorescence-based sensing instrument and method applicable to a broadrange of materials involving exponential decay and time delaymeasurements, and which is made from inexpensive components. Whileanalog systems of the present art are subject to drift and thereforeunnecessary errors, and the digital systems of the present art containcomplex, expensive hardware, the present invention has been designed toavoid these problems. Consequently, the system of the invention isinexpensive, convenient to use and operates over an extended andcontinuous measurement range. In addition, the measurement approach ofthe device and method of the invention is susceptible to convenient andprecise readout.

The foregoing description and the illustrative embodiments of thepresent invention have been described in detail in varying modificationsand alternate embodiments. It should be understood, however, that theforegoing description of the present invention is exemplary only, andthat the scope of the present invention is to be limited to the claimsas interpreted in view of the prior art. Moreover, the inventionillustratively disclosed herein suitably may be practiced in the absenceof any element which is not specifically disclosed herein.

I claim:
 1. An apparatus to measure emission time delay duringirradiation of targeted samples by utilizing digital signal processingto determine the emission phase shift caused by the sample, saidapparatus comprising:a source of electromagnetic radiation adapted toirradiate a target sample; means for generating first and second digitalinput signals of known frequencies with a known phase relationship;means for converting said first and second digital input signals toanalog sinusoidal signals; means for directing said first analog inputsignal to said electromagnetic radiation source to modulate saidelectromagnetic radiation source by the frequency thereof to irradiatesaid target sample and generate a target sample emission; means fordetecting said target sample emission and producing a correspondingfirst analog output signal having a phase shift relative to the phase ofsaid first analog input signal, said phase shift being caused by theemission time delay in said sample; means for producing a known phaseshift in said second analog input signal to create a second analogoutput signal; means for converting said first and second analog outputsignals to first and second digital output signals respectively; a mixerfor receiving said first and second digital output signals and comparingthe signal phase relationship therebetween to produce a signalindicative of the change in phase relationship between said first andsecond digital output signals caused by said target sample emission; andfeedback means to simultaneously alter the frequencies of said first andsecond digital input signals while substantially continuously vary thephase offset between said first and second digital input signals basedon said mixer signal to ultimately place said first and second digitaloutput signals in quadrature while compressing the frequency rangetherebetween.
 2. The apparatus as claimed in claim 1, wherein saidsignal generation means comprises a multiphase oscillator adapted togenerate said input signals at specified frequencies and specifiedphases in response to said mixer signal.
 3. The apparatus as claimed inclaim 2, wherein said feedback means comprises a long pass filter forextracting and amplifying said mixer signal, and wherein said multiphaseoscillator includes a frequency calculator for adjusting frequency ofsaid input signals and a phase calculator for adjusting the relativephase said variable input signals.
 4. The apparatus as claimed in claim3, wherein said mixer includes means for filtering out the sum frequencyof the first output and third input signals and the sum frequency of thesecond output and third input signals so that said analog-to-digitalconversion means digitizes only the said difference frequencies of theoutput signals and said mixer compares the phase of only the differencefrequency between the first output and third input signals with thedifference frequency between the second output and third input signals.5. The apparatus as claimed in claim 4, wherein said apparatus furthercomprises means for generating a fourth digital signal having afrequency the same as said second output signal, means for mixing saidsecond digital output signal with said fourth digital signal to create afeedback signal to said fourth signal generation means to adjust thephase of said fourth digital signal until it is in quadrature with saidsecond digital output signal, means for generating a fifth digitalsignal having a frequency substantially the same as said fourth digitalsignal, means for mixing said fifth digital signal with said firstdigital output signal and generating an adjustment output signaltherefrom, and means for varying the phase of said fifth digital signalbased on said adjustment output signal to place said fifth digitalsignal and said first digital output signal in quadrature to determinethe phase shift caused by the emission time delay of said targetedsample.
 6. The apparatus as claimed in claim 5, wherein said signalgeneration means comprises a multiphase oscillator having a frequencycalculator adapted to generate and adjust the frequencies of said firstand second input signals and a phase calculator to adjust the phase ofsaid second input signal in response to said mixer signal, and a secondsignal generation means to create said third input signal frequency, andwherein said feedback means comprises a low pass filter for extractingand amplifying said mixer signal and a second frequency calculator foradjusting the output of said second signal generation means.
 7. Theapparatus as claimed in claim 1, wherein said mixer comprises a phasedemodulator, and said feedback means includes a low pass filter forextracting and amplifying said phase demodulator signal and a phasecalculator for adjusting the phase of said second digital signal toultimately place said first and second output signals in quadrature. 8.The apparatus as claimed in claim 1, wherein said targeted samplecomprises a fluorescent sample exposed to a light source modulated bysaid first input signal to cause said sample to generate fluorescenceemissions having said phase shift.
 9. The apparatus as claimed in claim1, wherein said targeted sample comprises turbid media exposed to alight source modulated by said first input signal to cause said sampleto emit time delayed scattered radiation having said phase shift.
 10. Anapparatus to measure emission time delay during irradiation of targetedsamples by utilizing digital signal processing to determine the emissionphase shift caused by the sample, said apparatus comprising:a source ofelectromagnetic radiation adapted to irradiate a target sample; meansfor generating first and second digital input signals of knownfrequencies with a known phase offset; means for converting said firstand second digital input signals to analog sinusoidal signals; means fordirecting said first analog input signal to said electromagneticradiation source to modulate said electromagnetic radiation source bythe frequency thereof to irradiate said target sample and generate atarget sample emission; means for detecting said target sample emissionand producing a corresponding first analog output signal having a phaseshift relative to the phase of said first analog input signal, saidphase shift being caused by the emission time delay in said sample;means for producing a known phase shift in said second analog inputsignal to create a second analog output signal; means for convertingsaid first and second analog output signals to first and second digitaloutput signals respectively; a mixer for receiving said first and seconddigital output signals and comparing the signal phase relationshiptherebetween to produce a signal indicative of the change in phaserelationship between said first and second digital output signals causedby said target sample emission; and feedback means to simultaneouslyalter the frequencies of said first and second digital input signalswhile substantially continuously varying the phase offset between saidfirst and second digital input signals based on said mixer signal toultimately place said first and second digital output signals inquadrature while compressing the frequency range therebetween.
 11. Theapparatus as claimed in claim 10, wherein said apparatus furthercomprises means for directing said first and second output signalsindividually through an antialiasing filter prior to saidanalog-to-digital conversion means.
 12. The apparatus as claimed inclaim 10, wherein said signal generation means comprises a multiphaseoscillator adapted to generate said input signals at specifiedfrequencies and to adjust the relative phase of said generated inputsignals in response to said mixer signal.
 13. The apparatus as claimedin claim 12, wherein said feedback means further comprises a phasecalculator adapted to receive said mixer signal and determine relativephase of input signals and a frequency calculator adapted to receivesaid mixer signal and determine frequency of said input signals.
 14. Theapparatus as claimed in claim 10, wherein targeted sample comprises afluorescent sample exposed to a light source modulated by said firstinput signal to cause said sample to generate fluorescence emissionshaving said phase shift.
 15. The apparatus as claimed in claim 10,wherein said targeted sample comprises turbid media exposed to a lightsource modulated by said first input signal to cause said sample to emittime delayed scattered radiation having said phase shift.
 16. Theapparatus as claimed in claim 10, wherein said signal generation meansis adapted to generate a third input signal, and wherein said apparatusfurther comprises a first signal down conversion means positioned forcombining the frequency of said third input signal with the frequency ofsaid first output signal to produce a modified first output signalrepresenting the sum and difference frequencies of said first output andsaid third input signals, and a second signal down conversion meanspositioned for combining the frequency of said third input signal withthe frequency of said second output signal to produce a modified secondoutput signal representing the sum and difference frequencies of saidsecond output and said third input signals, said analog-to-digitalconversion means, converting said first and second modified outputsignals to digital signals for receipt by said mixer.
 17. An apparatusto measure emission time delay during irradiation of targeted samples byutilizing digital signal processing to determine the emission phaseshift caused by the sample, said apparatus comprising:a source ofelectromagnetic radiation adapted to irradiate a target sample; meansfor generating first, second and third digital input signals of knownfrequencies with known phase offsets; means for converting said first,second and third digital input signals to analog sinusoidal signals;means for directing said first analog input signal to saidelectromagnetic radiation source to modulate said electromagneticradiation source by the frequency thereof to irradiate said targetsample and generate a target sample emission; means for detecting saidtarget sample emission and producing a corresponding first analogintermediate signal having a phase shift relative to the phase of saidfirst analog input signal, said phase shift being caused by the emissiontime delay in said sample; first signal down conversion means forcombining the frequency of said third analog input signal with thefrequency of said first analog intermediate signal to produce a firstanalog output signal representing the sum and difference frequencies ofsaid first analog intermediate and said third analog input signals;means for producing a known phase shift in said second analog inputsignal to create a second analog intermediate signal; second signal downconversion means for combining the frequency of said third analog inputsignal with the frequency of said second analog intermediate signal toproduce a second analog output signal representing the sum anddifference frequencies of said second analog intermediate and said thirdanalog input signals; means for converting said first and second analogoutput signals to first and second digital output signals, respectively;a mixer for receiving said first and second digital output signals andcomparing the signal phase relationship therebetween to produce a signalindicative of the change in phase relationship between said first andsecond digital output signals caused by said target sample emission; andfeedback means to simultaneously alter the frequencies of said first andsecond digital input signals while substantially continuously varyingthe phase offset between said first and second digital input signalsbased on said mixer signal and to alter the frequency of said thirddigital input signal to achieve desired downconversion frequency of saidfirst and second digital output signals to ultimately place said firstand second digital output signals in quadrature while compressing thefrequency range therebetween.
 18. The apparatus as claimed in claim 17,wherein said mixer includes means for filtering out the sum frequency ofsaid first intermediate and third input signals and the sum frequency ofsaid second intermediate and third input signals to enable saidanalog-to-digital conversion means to digitize only the differencefrequencies of said first and second output signals, said mixercomparing the phase of only the difference frequency between said firstand second output signals.
 19. The apparatus as claimed in claim 18,wherein said apparatus further comprises means for generating a fourthdigital signal having a frequency the same as said second output signal,means for mixing said second digital output signal with said fourthdigital signal to create a feedback signal to said fourth signalgeneration means to adjust the phase of said fourth digital signal untilit is in quadrature with said second digital output signal, means forgenerating a fifth digital signal having a frequency substantially thesame as said fourth digital signal, means for mixing said fifth digitalsignal with said first digital output signal and generating anadjustment output signal therefrom, and means for varying the phase ofsaid fifth digital signal based on said adjustment output signal toplace said fifth digital signal and said first digital output signal inquadrature to determine the phase shift caused by the emission timedelay of said targeted sample.
 20. The apparatus as claimed in claim 18,wherein said signal generation means is adapted to create said first andsecond input signals with substantially the same frequencies and saidthird input signal with a substantially different frequency.
 21. Amethod for measuring emission time delay during the irradiation oftargeted samples by utilizing digital signal processing for determiningthe emission phase shift caused by irradiation of the sample, saidmethod comprising the steps of:generating first and second digital inputsignals of known frequencies having a known variable phase relationship;converting said first and second digital input signals to analogsinusoidal signals; directing said first analog input signal to anelectromagnetic radiation source to modulate the emissions of saidelectromagnetic radiation source by the frequency thereof; irradiating atarget sample with the modulated emissions of said electromagneticradiation source to generate a target sample emission; detecting saidtarget sample emission and producing a corresponding first analog outputsignal having a phase shift relative to the phase of said first analoginput signal, said phase shift being caused by the emission time delayof the emissions in said sample; producing a known phase shift in saidsecond analog input signal to create a second analog output signal;converting said first and second analog output signals to first andsecond digital output signals, respectively; mixing said first andsecond digital output signals and comparing the signal phaserelationship therebetween to produce an error signal indicative of thechange in phase relationship between said first and second digitaloutput signals caused by said target sample emission; and altering thefrequencies of said first and second digital input signals whilecontinuously varying the phase offset between said first and seconddigital input signals to ultimately place said first and second digitaloutput signals in quadrature while compressing the frequency rangetherebetween.
 22. The method as claimed in claim 21, wherein said methodfurther comprises the steps of generating a third input signal with saidfirst and second input signals, down converting said first analog outputsignal by combining the frequency of said third input signal with thefrequency of said first analog output signal to produce a modified firstoutput signal representing the sum and difference frequencies betweensaid first output and said third input signals, and down converting saidsecond analog output signal by combining the frequency of said thirdinput signal with the frequency of said second analog output signal toproduce a modified second output signal representing the sum anddifference frequencies between said second output and said third inputsignals, said first and second modified output signals being convertedto digital signals.
 23. The method as claimed in claim 22, wherein thesum frequency of the first output and third input signals and the sumfrequency of the second output and third input signals are filtered outso that only the difference frequency between the first output and thirdinput signals is mixed and phase compared with the difference frequencybetween the second output and third input signals.
 24. The method asclaimed in claim 22, wherein method further comprises the step ofgenerating a fourth digital signal having a frequency the same as saidsecond output signal, mixing said second digital output signal with saidfourth digital signal to create a feedback signal to said fourth signalgeneration means to adjust the phase of said fourth digital signal untilit is in quadrature with said second digital output signal, generating afifth digital signal having a frequency substantially the same as saidfourth digital signal, mixing said fifth digital signal with said firstdigital output signal to generate an adjustment output signal therefrom,and varying the phase of said fifth digital signal based on saidadjustment output signal to place said fifth digital signal and saidfirst digital output signal in quadrature to determine the phase shiftcaused by the irradiation of said targeted sample.
 25. A method formeasuring emission time delay during irradiation of targeted samples byutilizing digital signal processing to determine the emission phaseshift caused by the sample, said method comprising the stepsof:generating first, second and third digital input signals of knownfrequencies with known phase offsets; converting said first, second andthird digital input signals to analog sinusoidal signals; directing saidfirst analog input signal to an electromagnetic radiation source tomodulate said electromagnetic radiation source by the frequency thereof;irradiating said target sample with the modulated emissions of saidelectromagnetic radiation source to generate a target sample emission;detecting said target sample emission and producing a correspondingfirst analog intermediate signal having a phase shift relative to thephase of said first analog input signal, said phase shift being causedby the emission time delay in said sample; down converting said firstanalog intermediate signal by combining the frequency of said thirdanalog input signal with the frequency of said first analog intermediatesignal to produce a first analog output signal representing the sum anddifference frequencies of said first analog intermediate and said thirdanalog input signals; producing a known phase shift in said secondanalog input signal to create a second analog intermediate signal; downconverting said second analog intermediate signal by combining thefrequency of said third analog input signal with the frequency of saidsecond analog intermediate signal to produce a second analog outputsignal representing the sum and difference frequencies of said secondanalog intermediate and said third analog input signals; converting saidfirst and second analog output signals to first and second digitaloutput signals, respectively; mixing said first and second digitaloutput signals and comparing the signal phase relationship therebetweento produce a mixer signal indicative of the change in phase relationshipbetween said first and second digital output signals caused by saidtarget sample emission; and simultaneously altering the frequencies ofsaid first and second digital input signals while substantiallycontinuously varying the phase offset between said first and seconddigital input signals based on said mixer signal and to alter thefrequency of said third digital input signal to achieve desireddownconversion frequency of said first and second analog and digitaloutput signals to ultimately place said first and second digital outputsignals in quadrature while compressing the frequency rangetherebetween.
 26. An apparatus to measure emission time delay duringirradiation of targeted samples by utilizing digital signal processingto determine the emission phase shift caused by the sample, saidapparatus comprising:a source of electromagnetic radiation adapted toirradiate a target sample; means for generating first, second and thirddigital input signals of known frequencies with a known phaserelationship; means for converting said first, second and third digitalinput signals to analog sinusoidal signals; means for directing saidfirst analog input signal to said electromagnetic radiation source tomodulate said electromagnetic radiation source by the frequency thereofto irradiate said target sample and generate a target sample emission;means for detecting said target sample emission and producing acorresponding first analog output signal having a phase shift relativeto the phase of said first analog input signal, said phase shift beingcaused by the emission time delay in said sample; means for producing aknown phase shift in said second analog input signal to create a secondanalog output signal; a first signal down conversion means positionedfor combining the frequency of said third analog input signal with thefrequency of said first analog output signal to produce a modified firstanalog output signal representing the sum and difference frequencies ofsaid first analog output and said third analog input signals, and asecond signal down conversion means positioned for combining thefrequency of said third analog input signal with the frequency of saidsecond analog output signal to produce a modified second analog outputsignal representing the sum and difference frequencies of said secondanalog output and said third analog input signals; means for convertingsaid first and second modified analog output signals to first and seconddigital output signals, respectively; a mixer for receiving said firstand second digital output signals and comparing the signal phaserelationship therebetween to produce a signal indicative of the changein phase relationship between said first and second digital outputsignals caused by said target sample emission; and feedback means toalter the phase of the second digital input signal based on said mixersignal to ultimately place said first and second digital output signalsin quadrature.